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Sunday, October 22, 2017

Differential Amplifier Probe : Make, Buy or Both?


Why (I think) I Need a Differential Probe

After I sold my trusted Fluke 123 ScopeMeter, I lost the ability to connect probes where ever I wanted to, without thinking, or risking damage.

With my new -earth grounded- scope, I need to be extra careful to avoid making shorts by clipping the ground leads to the appropriate spots. Something I need to get used to again.

The answer, of course, is to use a Differential Probe as the front-end to an ordinary, ground or rather, an earth connected scope. As has been reported many times before, these instruments are ranging in price from expensive to outrageously expensive, even to several times the price of my 350MHz scope.


Buying Options

A new offering just came on the market, and that particular probe of 1000V @ 100MHz can be had for about $160.
Micsig Differential Probe
Unfortunately, there are no reviews that I could find yet. The specifications are very good for this price. Too good maybe?

The almost industry standard probe is available from many sources, and is made by one company.
Sapphire
It's about the same price, but only offers 25MHz, the 100MHz version is a lot more expensive.
Professional Differential Probe

Depending on the applications for a probe like this, it's important to list what you really want to do with it. In my case, I occasionally want to look at line level voltages, and for me that means 230V AC at 50Hz. For that 25MHz is plenty. I don't have 3-phase voltages, so 500V is plenty and safe enough. I'm a hobbyist, so official CAT III is nice to have but an overkill if you know what you're doing. (note the word kill)

Other applications I more regularly do is measuring in not ground connected circuits, like across a series shunt in a power supply. In that case, we're talking levels of a few milli-volt to a few volts, but the potential is most likely between 15-50V DC. Other applications are across collector/emitter or source/drain measurements, with the same voltage levels (<50V DC). In those cases, I would like to measure higher frequencies, together with their harmonics, or with a decent pulse/edge representation and in that case, having a bandwidth of 100MHz gives you the headroom that a 25MHz probe just can't give you.

For the majority of my applications, 50V @ 100MHz is plenty, so do I really need to spend $160 for a 1300V safety net, or can I build something myself?

When you look under the hood of these commercial probes, there is not a whole lot there that would justify the price.
Under the Hood
More details
Reverse-Engineered Schematics


DIY Investigation

If you look at the schematics, the value of the parts is not a lot, except for maybe the dual JFET, which can cost about $10-20. So the rest must be labor time in additional manual assembly and adjusting these probes during final test, and of course very healthy profit margins. If you look at the Sapphire 25MHz version probe circuit, apart from the attenuate section, what is standing out are the many R/C filter components and adjustments. If you look at the pictures of the board, there are several parts that look like they were added, tweaked or changed during the testing or calibration. It has all the looks of an untamed tiger to me.

For quite some time, I was on the fence between buying or making.
So, as a matter of interest, I spent quite some time trying to understand what the difficulty is in building a DIY version that will do what I need. Unfortunately, there is not a whole lot of DIY information on the Web, which surprised me a bit.

In essence, what you need is a differential input and a summing amplifier between the Device Under Test (DUT) and the scope. I found three different typologies that were described in various posts when I did my searching. I tried several of them out in LTspice to learn about the circuits and see if I could tailor them to my needs.

Here is a design that has been done for the automotive industry. I don't have the full paper, you need to buy that, but there is some interesting information that can be gathered from the pictures and tables.
Information is here : link
Among the viewable information is a comparison between the various commercial offerings:



The 3 Major Typologies

There may be more, but this is what I focused on. Note that for this topology section, I used one of my typical go-to prototype op amps, the TL072, which has a JFET input, wide supply voltage range and a Unity Gain B/W of 3MHz. This is just to show the concepts.

Simple single OpAmp design:


Very simple, if you don't need a lot of attenuation, because the frequency compensation becomes a bit cumbersome, and it's not so easy to get the required bandwidth. Using only one op amp to do everything has it's price. Here is an example circuit : Here
 The site is in Dutch, so you may have to use Google Translate.

Next is the more traditional Instrumentation Amplifier Design. You can build this using separate op-amps, or use a "real" differential op-amp (also called an instrumentation op-amp) like the AD8421.


Because this seems to be the most logical circuit to use, I spend quite some time trying to understand and to make this work for me. The good news is that it's easy to adjust the gain, by changing R8, the 100K resistor. The not so good news is that there are 6 resistors that need to have the same value (the 499 Ohm ones). Otherwise, the gain, or the CMRR is compromised. I found an application using this typology here : Example
And also here : Here  This one is in Dutch so you may have to use Google Translate. Here is another one: Here
Unfortunately, both projects were not completely finished but have a lot of excellent information.

The Gerber files for the first and third project are available, so these designs can be duplicated rather easily. Note that the Gerber files on the website for the first project are not complete and have errors. I contacted Kalle to see if he is willing to update the information. Kalle used the LT1818 and LT1819 (dual version) for his design, but there are some things you need be aware of when using these op amps. These op amps are relatively inexpensive with 2,53 Euro for the -18 and 5,17 Euro for the -19 version at Conrad.com.

Here is the first project as a circuit in LTsim:


As you can see, the simulation does not allow for more than 100MHz, still OK for my use. But this circuit needs some more work. See below.


And finally, Topology 3
This one is really straightforward.


I found this typology on a website here : diff amp probe

As with the previous design, Steven shared his design and Gerber files on OSH Park, so there are PCB's easily available. This is important, because I don't underestimate the layout challenges for 100MHz signals myself. The other thing I like is the utter lack of frequency compensation needed for the THS4631 he selected.
One down side is that this critter is rather expensive at 11,63 Euro (Conrad.com), and you need 3.

Here is the LTsim circuit with the THS4631's.


From the Bode plot, it is immediately apparent that although the op amp is specified for 325MHz when operated in Unity Gain, the summing amplifier action reduces this to about 108MHz @ -3dB. The Bode plot for the outputs of Unity Gain buffers U1 and U2 were indeed about 350MHz @ -3dB.

This simple circuit did not need any frequency compensation at all, which should not be underestimated.
Here is the Step Response with an input of 1Vp-p 50ns pulses with 100ps edges.


Looks great to me. To make this circuit usable, we now need to look at the maximum input and output voltages. The overall probe circuit is designed to have Unity Gain, so what goes in, also comes out at the same level.

Slew Rate
However, here is a hurdle that we need to keep in mind. The Slew Rate (SR) of an op amp determines the maximum voltage swing we can expect at a certain frequency. Normally this is no big deal, but when you're working at the limits of the bandwidth, this "rate of change" becomes an important factor.

The TI datasheet for the THS4631 has this:
High Slew Rate:
- 900 V/us (G=2)
- 1000 V/us (G=5)

There is no unity gain number listed. In any case, what this means is that with a 1V voltage swing (From 0 to 1V) at 100 MHz, we will need a minimum SR of :
     2 x Pi x 1V x 100MHz = 628 V/us

Or, in other words, with the THS4631, we can only go from 0 to 1.4V max. @ 100MHz.

The LT1818/19 has an SR of 2500, much better, but still not enough for 5V logic levels.

To circumvent that limitation, we will need attenuation for the input signals. If the input levels to the op-amps stay below 1V, we can use attenuation circuits for 10V, 100V and 1.000V and still have hope to keep close to the 100MHz bandwidth.

It's unclear how Steven works with his circuit without attenuation, maybe he uses a 10x or 100x scope probe. I have asked him for his comments on this, but I did not get a reply from him at all.


Input Attenuation

For my applications, it makes sense to have a 10V max input level for logic signals, 100V for higher voltages like power supplies etc., and 500V for mains related voltages. I'm going to design that last stage for 1.000V. This makes the math easier when you look at the scope signals, because the attenuation will be 10x (-20dB), 100x (-40dB) and 1.000x (-60dB). This will allow me to have a maximum input of 1Volt and also an and output of the circuit of 1V going to the scope, and that is well below the maximum SR value.

In principle, the attenuation circuit is a combination of a resistor divider for the lower frequencies and a capacitor divider for the higher frequencies. The simple version looks like this:


Before I started to investigate differential probes, I did not yet know the seemingly magic relationship between the above component values. Apparently, for a flat frequency response from DC all the way up to the (in this case) 100MHz bandwidth, the two capacitors have to have the inverse ratio as the two resistors have.  Initially, I was surprised to see several input circuits from very different designs that all came down to this basic circuit. I tried a few variations in LTsim and they all worked. I learned something new again.

This basic circuit can be easily changed for 1.000x or 10x attenuation by changing the resistor values and keeping the capacitor values with the same ratio.
To calculate C3, divide R1 by R2, and multiply that (ratio) result times the value of C2

         C3 = R1 / R2 * C2

In the above circuit, the DUT will see a loading impedance of just over 1M Ohm (1M + 10K). This is the equivalent of a passive 1x oscilloscope probe, which still may be too high of a load on the DUT and could even influence the operation of the DUT, causing measurement errors. The Sapphire probes (the 9001 version) use 4 Meg and 26K dividers, resulting in a 0.65 attenuation factor, and this result gets amplified later on. I would like to keep Unity Gain from input to output, so that limits me to decade numbers to make the math easy.

The higher impedance is more important for me in the 10V and 100V probe versions, due to the circuits I will probe. In this case, going to the equivalent of a 10x scope probe (10Meg impedance) is desired. This higher impedance can be accomplished by using a series resistor of 9Meg together with a 1M resistor, resulting in a 10Meg impedance to the DUT. Alternatively, you could go half way, by using a series resistor of  4 or 5 Meg.

For the 1.000V version, it's not so much the loading of the circuit, but much more the safety aspect that plays an important role. So in this case, you could still use the 9M resistor, but combined with a 10K instead of the 1Meg, resulting in a x1.000 (-60dB) attenuation. The 9Meg resistor MUST be a high voltage version, or otherwise a composition of high voltage resistors in series that create enough of a spark gap and creeping distance between the DUT input and the rest of the probe circuits.

At this moment, and for the simulation phase, I'm using the following values:
For the LT1818:
10x (-20dB)     1M || 12 pF and 111K111  || 108pF
* See Measurement error below

For the THS4631:
10x  (-20dB)     9M || 12 pF and 1M || 108pF

For both:
100x  (-40dB)   9M || 12 pF and 90K909 || 1.188nF
1.000x (-60dB) 9M || 12 pF and 9K009 || 11.988nF


Measurement Errors

Due to the high input resistor value of 9Meg, there will be a measurement error caused by the 1.5nA input bias current of the THS4631 buffer amps. This is acceptable to me, even with the -20dB attenuation factor. The LT1818/19 however have a 10uA bias current, which is a lot higher! With a -60dB attenuation and a 1.000 Volt input, this current will cause an error of 90V over the 9M resistor. For me, that error even with the -20dB attenuation is too large. When that is a concern, you need to lower the resistor values. In many cases however, precision at this voltage level is probably not needed, and loading of the DUT is more important. An interesting dilemma.

Another error source can be caused by the protection diodes. Kalle used BAV99 diodes. Their added capacitance to the input is only 1.5pF each, which is very good, but the reverse current per diode is 2.5uA, adding 5uA to the error budget for each input channel. At a 1.000 Volt at the input, this causes a voltage drop over the 9M resistor of another 45V! Keep those error conditions and causes in mind.

I suggest you use the 1N4148 in the SMD package. Their added capacitance is a little higher at 4pF, but the reverse current is only 25nA resulting in a voltage drop of only 0.2V.

When I have a running version of the amplifier section, I will prototype and decide on the final input resistor values that I will use.


Common-Mode Rejection Ratio (CMRR)

Apart from measuring signals while not connecting one probe connection to earth or ground, is to measure the voltage difference between any two points in a circuit. The added benefit is that any coupled noise that is entering the two measurement leads, get subtracted out, so you can see and measure the pure signal. For our differential probe, we need to take some steps in order to keep the CMRR as high as possible. This means that the resistance and capacitance that are presented at the input of the input buffers and the summing amplifier need to be exactly the same. This challenge is probably the most important reason for the price justification for the commercial probes. The THS4631 lists a CMRR of 80-95dB, so that should be the goal for the overall probe.

To allow for inevitable component tolerances, we need to make the attenuation circuit somewhat adjustable to make both input connections (arms) equal.

If you look at the various circuits in the above examples, they all boil down to an input attenuation section that looks very much like this:



Both the 12pF capacitors can be constructed by putting 4 x 47pF capacitors in series. This will even out the tolerances, and will create a "spark" gap between the high voltage input and the 1V level output of this circuit. Any remainder high frequency adjustment can be made at the summing op amp, to account for any other capacity issues on the PCB, I hope.

The resister divider of both inputs must be equal in value, so a small 100-500 Ohm trimmer, here shown with R15 and R16 as if in the mid-point position, will allow to tweak that. The size of the trimmer will be dictated by the level of matching you can do on the resistors in the first place. The precision is less important, their identical values is. The 108pF section is a composite of a few capacitors in parallel. A capacitor trimmer will allow you to set the precise level, and that can be seen and adjusted at the output of the amplifier with a fast rise pulse.


Input Protection

If I can, I want to add some form of input protection for the expensive op amps, so I'll try to add a set of 1N4148 diodes to the voltage lines so they will clamp the inputs. The standard 1N4148 diode only adds 4pF, which is better than the typical 10-12 pF for a TVS diode. Besides, the 1N4148 is very inexpensive.

Below is the complete circuit diagram with the 10x attenuation (-20dB) using Steve's design. This can also be the base architecture for the x100 and the x1000 versions.


My hope is that I can use the amplifier circuit boards that are available through OSH Park, the Version 2 model, and add the three different front-end attenuate sections to them. Below are pictures from Steve's site:


At this moment, I'm pretty confident that I can make this work.


Reality Check

Now, just for a reality check:
What will the DIY make method cost me? Let's use Steven's design for this.

The 3 PCB's (minimum qty) from OSH Park will cost me $ 3.70 incl. shipping.
3 x THS 4631 will cost me 11,63 a piece, so about 35 Euros without shipping.
The rest of the parts is small change, say 10 Euro incl trimmers and SMA connector and maybe another 10-15 Euro for a case.

That will set me back at least 60 Euro's for one probe, so I can possibly get 3 probes for the price of one Micsig, but I don't get the test leads and adapters with it, nor do I have the means to really test the 1.000x version for the high voltages.

If I go the LT1818/19 way with Kalle's design, the chips are lower cost (only 8 Euro's per board), but due to the loading issues, there could be more at stake with these chips that I have not uncovered yet. The PCB has more real estate, so will be a bit more expensive. All in all, this route will still be less expensive, possibly between 20-25 Euro's less.

If I'm really brave, and consider making my own PCB, I will combine the two circuits. I will then use the THS 4631 for the input buffers, and the LT1819 for the summing amplifier. I tested that combination with LTsim and that works fine.

Hmmm, I need to think a bit more about the make or buy decision before I pull the trigger either way...

Update 31-oct-17:
I pulled the trigger on the THS4631 version, and ordered the PCB's, the chips (I found a source with 10 of them for about 43 Euros) and ordered enough of the 499R and 49.9R resistors to find some matched sets.

CMRR Simulation
 While continuing to learn and play with this project, I also learned how to do a CMRR simulation.


I'm not an expert, but this looks pretty good to me. A possible caveat is that LTspice may not do CMR simulations very well.

I also found a couple of hints and tricks to do a better analysis by using a Monte Carlo model for the most critical parts that have the greatest influence on the CMR. Obviously, they are the resistors and trimming capacitors for the attenuation input, and then the two resistors for the summing amp. The other components did not make a sizeable difference, so I left them at their previous fixed values. I used 1% for the attenuation resistors, 5% for the 12pF, 20% for the trimmer caps, and 1% for the two 499Ohm resistors for the summing amp.

These two last resistors largely define the CMR for the op-amp stage. I will need to carefully select them for a matched set out of the batch that I ordered.

Still looks pretty good to me with an average of about 84dB, and a worst case of 72dB. My goal was to get close to the 80-95dB set by the op-amps.



It will take a few weeks for the parts to arrive, but when I have the boards and populated one, I will start to work on the attenuation values.

Update 2-Nov-2017:
And then..... you get an email from Micsig with a special offer; two probes for the price of one!:
Micsig Combo offer
The offer is not valid anymore, but at the time, I almost pulled the trigger...
The current prices (aug 2019) for the Micsig DP10013 are now 170 Euros (incl. VAT).

The DP10013's are listed as having a spec of 1.300V and 100MHz, but that seems highly improbable with their front-end probing leads. I don't really care, because when I probe at line levels, a minimum of 25MHz is most likely enough, and the safety aspect is much more important for me. The DP10013 50x and 500x attenuation levels are more tailored to higher voltages and line level measurements.

In the meantime, the parts arrived and I completed the build of the three boards. A quick initial test showed that they all work fine. I need to finish a few other projects before I start to work on the attenuation circuits and do more measurements.

Update august 2019:
After I started this post and project, I left it alone for just almost two years. There were other projects that were more importent, and I did not have the tools I wanted/needed to really finish the probes. As an example, I did not have a fast rise pulse or a function generator that would go beyond 100MHz. Since then, I finished my project that would give me the fast rise pulse, so I can finally test and adjust the high frequency responses. I also moved away from Eagle and learned to work with Diptrace, including producing PCB's. And then, I also purchased the equipment to handle SMD parts. All these elements and steps were required to continue with this project.

I'm definitely NOT buying!
Interestingly, during this whole period, I never had the urging need to use a differential probe. This made me realize that buying a Micsig kit is not justified for my typical applications.

I'm making!
I have just ordered the parts for the attenuation sections, so when I have them, I can start to work on the PCB's. In the meantime I will prototype the front-ends and start some measurements.

Update Sept 2019
Believe it or not, but I have finally started again on this project.
Now that I have some better equipment, I can start to profile the amplifier boards.
At first I was a little disappointed when I saw the first results of the fast edge pulse response.
I made sure to stay below 1Vp-p to avoid slew-rate issues, but still...
Here is the screen shot:

Don't you love that fast rise pulse from the Tektronix PG506? I do!

To fix the hf response, I kludged a small 0-10pF variable capacitor with tiny strands to the very tiny pads across the output amplifier feed-back resistor, and tuned it for the optimum pulse response. Things started to look a lot better already. I measured the trim capacitor to be about 3pF, so that is the value that I soldered in place. Steve used a 1pF value on his board, but the Texas Instruments datasheet lists 5.2pF for this configuration. Just for reference. 

Here is the result of that tweak:


Could be a bit better, maybe I should try another value, but I did not want to fiddle too much with this at the moment, in fear of damaging the tiny pads on the pcb. I'll try a few other values on one of the other amplifiers boards. Update: I tried various other values, also with the other two boards but I'm not getting a much improved situation. I'm using 3pF for now.

You probably notice the rise time difference and phase shift. The fast rise pulse response of the amplifier is 3.3nS. Using the Bandwidth formula with the 0.4 factor, this should theoretically be 133MHz. Not bad at all.
To reiterate what I mentioned earlier, I can't verify the bandwidth with a sine-wave, because my FY6600 function generator is the 30MHz version.

Next step is to build a prototype 10x (-20 dB) attenuation board and have a look at what that does.

Below is my first attempt, it follows the schematic diagram of the -20dB version above, but without the protection diodes:


I have used 3 x 3Meg resistors in series to create a 9M resistor. Parallel to that string is a set of capacitors to get to the required 12pF. This is repeated for each input. There are 2 x 47Pf capacitors in parallel with the 40 pF trimmers. The terminating resistors are 1Meg and they are tied to ground with a 10K trimmer. I used 10K because the resistor values of the 9M and 1M are less than perfect, so needed much more tuning than the 100Ohm trimmer I put in the diagram above.

The -20dB attenuation is within a few percent at low frequencies, as can be expected. Unfortunately, with higher frequencies (above 500KHz)  I am getting oscillations.  I need to rethink the way I'm building this front-end and maybe go to a pcb layout already, something I wanted to avoid just yet. I don't know enough and have zero real experience to come up with a good pcb layout and frequently turning boards is what I want to avoid.

I have designed a pcb for the front-end, and sent it out to production. Here are the pictures of the layout. (I'm using DipTrace)



I put in a few extra places for components so I can fool around with this a bit. This is a test version that will allow me to setup different attenuation factors.

In the meantime, my long awaited nanoVNA (Vector Network Analyzer) tool arrived, and after some playing around with it, I decided to give it something to chew on. I created a calibration of the unit using the connectors and cables I was going to use, and then connected the amplifier.

The amplifier input is connected to CH0 (stimulus) and the output to CH1 (response) of the VNA. I set the frequency span to start from the minimum at 50KHz to 250MHz. Here is a picture of that setup and the result:


The two leads going to the input of the amplifier are supposed to mimic the flying leads I'm going to use later with mini grabbers to the DUT, at least that is my hope. I first calibrated the nanoVNA with this cable attached without the amplifier and used the typical open-short-load (50 Ohm) at the end of the cable. The output of the amplifier is 50 Ohm, so that should match with the input.

(note that in the lower left hand corner of the sheet, I scribbled notes while trying to use the NanoVNA to measure the bandwidth of my scope. It passes the 300MHz easily. One point to note is that the NanoVNA uses tricks to get beyond 300MHz, and to compensate, it increases the output volume, making it more complicated to find the real -3dB cutoff point for the scope)

Here is a screenshot of the PC program that talks to the NanoVNA, and allows me to use my normal glasses. The screen is very small, and the type fonts extremely tiny, not helping with my deteriorating eyesight.


I got this display after fiddling a bit with the curve of the input lead. This result looks a lot cleaner. What I think we can see here is that the output of the amplifier has a small 6dB rise at around 60MHz, and then stays just about flat until the -3dB point at about 145 MHz.
Not bad at all I think! Now I need to find a way to see if I can get rid of the 60 MHz issue.

Just when I was about to disconnect everything and call it a day, I noticed that the NanoVNA was still working and collecting data, even though I already pulled the batteries from the amplifier. I noticed that without power, there is the same hump at 60 MHz as we saw on the powered unit. No idea what I'm seeing here...yet



Stay tuned for more...




Saturday, October 14, 2017

Setting up an FFT Measurement System

In this post I'll describe how I use a number of components to create a system that you can use to measure harmonic distortion and noise by using FFT's. This system can be configured in several ways to verify and measure the distortion of amplifiers, sine wave generators, power supplies etc.

To measure the performance of a black-box system (the Device Under Test - DUT), a very clean and low distortion sine wave (the fundamental) is used to stimulate the DUT, while the output is sampled. The output signal is stripped from the fundamental sine wave by means of a sharp filter, and the remaining residual, the noise and distortion is fed to an FFT system that can show the results.

Following is a picture of the setup needed to measure harmonic and noise (DHT and DHT+N) from a DUT.






This is a setup that can be used to measure sine wave generators, or to verify the Sound Card interface to the PC and the FFT software.

A high quality low distortion 1KHz sine wave is fed to the DUT, here symbolized as an amplifier. The output of the DUT is fed to the Soundcard Interface and attenuated. This interface protects the delicate and sensitive inputs of the USB Sound Card, and also facilitates the various measurements. The resulting output is then fed to a Twin-T filter which removes the 1KHz fundamental. The residual (everything added to the pure sine wave) is then fed back into the Soundcard Interface and then going to the USB Sound Card Interface, where it is digitized and sent to the PC over the USB link. On the PC, the AudioTester software is used to show the residual (and noise) by means of an FFT.

This same setup can also be used to measure the noise performance of a power supply. In that case, the power supply output is first stripped from the DC component by using a capacitor (not shown), and then fed to the Sound Card Interface, and from there to the USB Sound Card for digitizing and on to the PC and the AudioTester FFT software.

If you want to measure the quality of sine wave generators of verify the components in the link, you can also use the following much simpler setup.



Here, the output of the sine wave generator is fed directly to the Twin-T filter and the residual is digitized by the USB Sound Card. The Sound Card Interface is not needed, because the input levels are low.




In an earlier post on this Blog, I already described how I put the the high quality sine wave generator, and an active Twin-T filter in an enclosure.




This instrument is described here:  simple but precise 1khz distortion system


I use a Sound Card Interface, to attenuate (high) signals coming from the DUT, which would otherwise destroy the input of the USB Sound Card, or the other measurement components. The output from the Interface then goes to the Twin-T filter and then to an actual USB Sound Card connected to a PC. The digitized output of the Sound Card is used by the AudioTester software running on the PC to show the results, typically by an FFT diagram.






The Sound Card Interface above is a DIY project and described here

During my first baby steps in putting this system together and collect some experience, I used this inexpensive (around 25 Euros) USB Sound Card Interface:


It worked OK as you can see in the posts I mentioned earlier, but I was not very impressed with the results. It needed further tweaking, adjusting and modifying. Since then, I moved on to other projects, so for a few years, I really didn't need this setup.

When I recently wanted to profile my DIY Tek SG502, I put the system together again and quickly made some changes that makes the connections between the units easier.

However, problems showed-up I had not seen before. It was probably caused partially because in the meantime, I switched to a newer and different Laptop and also upgraded to W10. I also could have done something wrong during some of my experiments, because I suspect that something in this USB Sound Card box could be damaged because it now shows a lot more harmonic distortion then I remember having seen before.

At first I was mystified to the cause, and could not put my finger on it. I now attribute it to a combination of my W10 Laptop, the W10 sound drivers, the W10 drivers for the USB Sound Card, plus something wrong with the drivers of the AudioTester software, because it is now crashing all the time.

Too many variables, so I started to address them one by one.
When I used my FY6600 DDS Function Generator, it showed THD+noise performance that looked pretty good and a little worse than the specifications. My unprofiled and just finished Tek SG502 was also just outside the specifications, but that was to be expected. However, the ultra pure 1KHz since wave generator from Victor showed results that were only a little bit better than the other two, so I started to suspect my reference.

But first, I wanted to buy a better USB Sound Card. I searched around, and found one that seemed to have the right performance, so I ordered an ASUS 

With the excellent help from Victor Mickevich, we found out by making some measurements, that the problem was not due to his sine wave generator.

I have a license for the AudioTester software, but I was not happy with the overall driver situation, and had problems with the calibration. To eliminate that aspect, I tried another software package, ARTA. I selected this, because there are many references on the web and examples of measurements made using Victor's oscillator, and the ARTA software, so I could start to compare. However, it did not improve on the root cause, the excessive harmonic distortion on Victor's oscillator.

It was now time to take the next step, and invest in a better USB Sound Card. I purchased a (brand new) ASUS Xonar U7. To my utter dismay, I found out that the CD that came with the software, did not have W10 drivers. C'mon ASUS, we're in the middle of 2017, and you have not updated the CD yet? Obviously, that did not give me a lot of confidence. Making a few measurements did not improve the situation much, so I returned the unit the next day.

Time for a reset. I realized that I lacked the knowledge to get to the bottom of this issue, so I had to learn a lot more first of all. I literally spend a few weeks going through all kinds of Forums and Blogs to see what other people were using, and to learn more about the overall system and the components in much more details.

Eventually, after going through many blogs and forum's, there were a number of USB Sound Cards that stood out. The majority were from the same company, the E-MU 0202, the E-MU 0404 and a few others, and, they were available on eBay now and then.
I decided to try to score one of these and after some miss-hits, I scored a used E-MU 0202.

The ultra-pure 1KHz sine wave generator I have, was designed by Victor Vickenich (vicnic), and he also published a couple of modifications to this Sound Card, with lots of pictures together with the FFT results using the same 1KHz source.

At this moment, we are out of the country for 7 weeks, so I will not get my hands on the 0202. This gave me the time to do some more investigations, so I started to study what Victor and others had done to the 0202 to get these incredibly good results. I took me a little while to find information, but I eventually found a few sources of  diagrams that together with his photo's of the PCB, allowed me to reconstruct the modifications. It turned out that the various sources of 0202 schematics I found had errors, or were incomplete.

Since I had nothing else to do at the moment, I took the time to capture all information in my schematic capture program (DipTrace), so I would have a record of the original status, and could put a description together for the modifications Victor, and maybe others did.

Note:
Afterwards, I found that the schematics for the 0202 are very similar to the 0404, although in a different layout and with several value changes and part numbering changes. Schematics for the 0404 are available, but I did not find good ones for the 0202.

Based on the various sources and the photographs of the PCB, here is the resulting schematic of the E-MU 0202 for the "B" channel input to the ADC.

E-MU 0202 Front-End Schematic Diagram Channel B 



And here is the more complex "A" channel:

E-MU 0202 Front-End Schematic Diagram Channel A


Note: Because I do not have the 0202 myself yet, I used photographs of the PCB and various sources I found, but I could not verify this for correctness. Use with care!

For the time being, I left out all power related parts. I'll add them when needed, because I'm considering adding a separate power input, instead of using the USB 5V from the PC. That's a potential project for later.

First of all, as soon as I have been able to verify the unit, I will start with the modifications that Victor made to his 0202. He eliminated much of the front-end of the unit. He uses an attenuation of his own, so there was no need for the input section. I will be using my Sound Card Interface if needed, so this is no limitation for me either. Cutting that input section out of the loop saves a number of dB's in noise, and turns the Sound Card more into a tailored measurement instrument.

Update!
I scored a EMU0202 on fleabay, and although it was supposed to be working, the output amplifier did not. I could not find the error, but I really didn't care. I wanted the digitizing front-end, so I applied the same modifications Victor published and did on his 0202.

Here are the two links to Victor's modifications and measurement results:
Modifications with results
Scroll down on the next page to get to the photographs with the modifications:
Detailed photo's of the mods
Note that a little below this post is another one with a correction to the value of R46, which needs to be 6K8.

Below is the schematic information I put together for the "B" channel, and the parts to be removed to isolate the front-end circuit to ADC input.




The volume control potmeter is removed to make place for an RCA or BNC connector on the front panel. I used a BNC connector myself.
The two removed series resistors, R54 and R32, both 1K4, will isolate the front-end input from the drivers for the ADC. The two resistors that are used to create the dynamic zero balance, R35 and R41 need to be removed too.

Following are the value changes and the new additions to form the new input circuit to the ADC.


The RCA or BNC connector can be mounted on the front panel in the hole of the potmeter. I had a BNC connecter this fitted perfectly. After that, the input series resistor, the capacitor and the input Z resistor can be mounted Manhattan style. Note that Victor mentioned that this resistor can be tweaked in value to remove artifacts. I kept mine to 220K.

Three feed-back resistors change in value.
R28 : replace the 1K value to 6K8.
R46 : replace the 1K value to 1K5
R34 goes from 1M to 1K. (Victor used the original removed 1K resistor from R28 and simply soldered that on top of R34 with the 1M value.)

Finally, the connection from the output of U6-B to the input of U6-A can be created by soldering the second removed 1K4 resistor, (the originals are too small, I used a new 1K5 0603) "Tomb Stone" and with a small wire to the input of U6-A. His detailed photographs show the way.

The result on my modified EMU0202 is rather stunning I think:


So my new EMU0202 digitizing front-end together with Victor's quality oscillator, this is a perfect reference combination for my applications.

Here is a screenshot of the EMU0202 with my DIY Tektronix SG502:


And here with my FeelTech FY6600-30 Dual Channel Function/Arbitrary Waveform Generator:



So with this setup I finally have a good starting point.
Next step will be to add in the Pete Miller Soundcard Interface and do some more measurements.

It may take a while, I have a few other projects I'm working on, but stay tuned for more...

Enjoy!